DTV broadcasting in the United States of America has been done in accordance with broadcasting standards formulated by an industry consortium called the Advanced Television Systems Committee (ATSC). ATSC published a Digital Television Standard in 1995 that employed 8-level vestigial-sideband amplitude modulation of a single radio-frequency (RF) carrier wave. This DTV transmission system is referred to as 8-VSB. In the beginning years of the twenty-first century efforts were made to provide for more robust transmission of data over broadcast DTV channels without unduly disrupting the operation of so-called “legacy” DTV receivers already in the field. These efforts culminated in an ATSC standard directed to broadcasting digital data for reception by mobile receivers being adopted on 15 Oct. 2009. This subsequent standard also used 8-level vestigial-sideband amplitude modulation of a single RF carrier wave, so the more robust transmission of data could be time-division multiplexed with the transmission of DTV signal to so-called “legacy” DTV receivers already in the field. The digital data to be transmitted for reception by mobile receivers employ Internet Protocol (IP) and are subjected to serial concatenated convolutional coding (SCCC) before being encapsulated in MPEG-2 transport stream (TS) packets. These TS packets are subsequently time-division multiplexed with the MPEG-2 TS packets of data transmitted for reception by “legacy” DTV receivers.
DTV broadcasting in Europe has employed coded orthogonal frequency-division multiplexing (COFDM) that employs a multiplicity of RF carrier waves closely spaced across each 8-MHz-wide television channel, rather than a single RF carrier wave per television channel. Adjacent carrier waves are orthogonal to each other. Successive multi-bit symbols are selected from a serial data stream and used to modulate respective ones of the multiplicity of RF carrier waves in turn, in accordance with a conventional modulation scheme—such as quaternary phase shift keying (QPSK) or quadrature amplitude modulation (QAM). QPSK is preferably DQPSK, using differential modulation that is inherently insensitive to slowly changing amplitude and phase distortion. DPSK simplifies carrier recovery in the receiver. Customarily, the QAM is either 16QAM or 64QAM using square 2-dimensional modulation constellations. In actual practice, the RF carrier waves are not modulated individually. Rather, a single carrier wave is modulated at high symbol rate using QPSK or QAM. The resulting modulated carrier wave is then transformed in an inverse discrete Fourier transform (I-DFT) procedure to generate the multiplicity of RF carrier waves each modulated at low symbol rate.
In Europe, broadcasting to handheld receivers is done using a system referred to as DVB-H. DVB-H (Digital Video Broadcasting—Handheld) is a digital broadcast standard for the transmission of broadcast content to handheld receivers, published in 2004 by the European Telecommunications Standards Institute (ETSI) and identified as EN 302304. DVB-H, as a transmission standard, specifies the physical layer as well as the elements of the lowest protocol layers. It uses a power-saving technique based on the time-multiplexed transmission of different services. The technique, called “time slicing”, allows substantial saving of battery power. Time slicing allows soft hand-over as the receiver moves from network cell to network cell. The relatively long power-save periods may be used to search for channels in neighboring radio cells offering the selected service. Accordingly, at the border between two cells, a channel hand-over can be performed that is imperceptible by the user. Both the monitoring of the services in adjacent cells and the reception of the selected service data can utilize the same receiver front end.
In contrast to other DVB transmission systems, which are based on the DVB Transport Stream adopted from the MPEG-2 standard, the DVB-H system is based on Internet Protocol (IP). The DVB-H baseband interface is an IP interface allowing the DVB-H system to be combined with other IP-based networks. Even so, the MPEG-2 transport stream is still used by the base layer. The IP data are embedded into the transport stream using Multi-Protocol Encapsulation (MPE), an adaptation protocol defined in the DVB Data Broadcast Specification. At the MPE level, DVB-H employs an additional stage of forward error correction called MPE-FEC, which is essentially (255, 191) transverse Reed-Solomon (TRS) coding. The transverse direction is orthogonal to the direction of the (204, 188) Reed-Solomon (RS) coding employed both in DVB-H and in DVB-T terrestrial broadcasting to stationary DTV receivers. This TRS coding is reported to reduce the S/N requirements for reception by a handheld device by a 7 dB margin compared to DVB-T. The block interleaver used for the TRS coding engenders a specific frame structure, called the “FEC frame”, for incorporating the incoming data of the DVB-H codec.
The physical radio transmission of DVB-H is performed according to the DVB-T standard and employs OFDM multi-carrier modulation. DVB-T employed coded orthogonal frequency-division multiplexing (COFDM) in which an 8 MHz-wide radio-frequency (RF) channel comprises 2000 or 8000 evenly-spaced carriers for transmitting to stationary receivers. DVB-T2, an upgrade of DVB-T proposed in 2011, further permits approximately 4000 evenly-spaced carrier waves better to accommodate transmitting to mobile and handheld receivers. These choices as to number of carrier waves are commonly referred to as 2K, 8K and 4K options. DVB-H uses only a fraction—e.g., one quarter—of the digital payload capacity of the RF channel.
DVB-T2 employs low-density parity check (LDPC) coding as forward-error-correction (FEC), rather than using concatenated convolutional coding (CCC) or using product coding. An LDPC code is based on an H matrix containing a low count of ones. Encoding uses equations derived from the H matrix to generate the parity check bits. Decoding is accomplished using these equations with “soft-decisions” as to transmitted symbols to generate new estimates of the transmitted symbols. This process is repeated in an iterative manner resulting in a very powerful decoder. Like parallel concatenated convolutional coding (PCCC), LDPC codes are subject to error floors. Outer coding, such as Bose-Chaudhuri-Hocquenghem (BCH) coding, can be added to LDPC technology to overcome error floor phenomena. LDPC coding provides AWGN performance that can approach the Shannon Limit even more closely than PCCC.
COFDM may again be considered for DTV broadcasting in the United States of America, where 6-MHz-wide rather than 8-MHz-wide RF channels are employed for such broadcasting. Generally, the 2K, 8K and 4K options are retained in proposals for such DTV broadcasting, with bit rates being scaled back to suit the 6-MHz-wide RF channels.
COFDM is able to overcome frequency-selective fading quite well, but reception will fail when there is severe flat-spectrum fading. Such flat-spectrum fading is sometimes referred to as a “drop-out” in received signal strength. Such drop-out occurs when the receiving site changes such that a sole effective signal transmission path is blocked by an intervening hill or structure, for example. Because the signaling rate in the individual OFDM carriers is very low, COFDM receivers are capable of maintaining reception despite drop-outs that are only a fraction of a second in duration. However, drop-outs that last as long as a few seconds disrupt television reception perceptibly. Such protracted drop-outs are encountered in a vehicular receiver when the vehicle passes through a tunnel, for example. By way of further example of a protracted drop-out in reception, a stationary DTV receiver may briefly discontinue COFDM reception when receiver synchronization is momentarily lost during dynamic multipath reception conditions, such as caused by aircraft flying over the reception site.
The ATSC standard directed to broadcasting digital television and digital data to M/H receivers used TRS coding that extended over eighty dispersed-in-time short time-slot intervals, rather than being confined to a single longer time-slot interval. A principal purpose of the TRS coding that extended over eighty time-slot intervals was overcoming occasional protracted drop-outs in received signal strength. The DVB-SH standard that will replace the DVB-H standard also employs long-duration TRS coding that extends over a few data frames. Confining TRS coding to a single longer time-slot interval as done in DVB-H is advantageous, however, in that error-correction is completed within a shorter time. This helps speed up changes in RF-channel tuning, for example.
Iterative-diversity transmissions were proposed to ATSC to facilitate alternative or additional techniques for dealing with flat-spectrum fading of 8-VSB signals. Some of these proposals were directed to separate procedures being used for decoding earlier and later transmissions of the same coded data to generate respective sets of data packets, each identified after such decoding either as being probably correct or probably incorrect. Corresponding data packets from the two sets were compared, and a further set of data packets was chosen from the ones of the compared data packets more likely to be correct. A. L. R. Limberg proposed delaying earlier transmissions of concatenated convolutionally coded (CCC) data so as to be concurrently available with later transmissions of similar CCC data, then decoding the contemporaneous CCC data with respective turbo decoders that exchanged information concerning soft data bits to secure coding gain. These various iterative-diversity transmission techniques, although comparatively robust in regard to overcoming additive White Gaussian noise (AWGN), were not incorporated into the ATSC 8-VSB system for DTV broadcasting since supposedly the single-time retransmissions halved available digital payload.
Iterative-diversity reception implemented at the transfer-stream (TS) data-packet level does not require as much delay memory for the earlier transmitted data as delaying complete earlier transmissions to be concurrent with later transmissions of the same data. This is because the redundant parity bits associated with FEC coding contained in those complete earlier transmissions is removed during its decoding and so do not need to be delayed. However, implementation of diversity reception at the TS data-packet level sacrifices the substantial coding gain that can be achieved by decoding delayed earlier transmissions concurrently with later transmissions of similar data and interchanging preliminary decoding results between the concurrent decoding procedures. Implementation of diversity reception at the TS data-packet level is also incompatible with code-combining of delayed earlier transmissions and later transmissions of similar data being used to improve signal-to-noise ratio (SNR).
With developing memory technology, it is becoming feasible to delay complete earlier transmissions of DTV time-slices for a few seconds in physically small memory that consumes little power and is practical for inclusion in a handheld receiver. Developments that will be commercialized in just a few years will make it feasible to delay transmissions of entire DTV time-slices for several seconds within an M/H receiver. Such delay memory is required for implementing long-duration TRS coding or for implementing iterative-diversity reception of once-repeated transmissions, whichever is done. Memory capable of delaying the initial ones of once-repeated transmissions for a number of seconds allows a DTV receiver to overcome severe drop-outs in received signal strength substantially as long as that number of seconds. If the same memory were used to support TRS coding over that number of seconds, the capability of a DTV receiver to overcome severe drop-outs in received signal strength is only a fraction of that number of seconds.
If long-duration TRS coding is used in COFDM transmissions of DTV, the DTV receiver will usually include an OFDM demodulator followed by employ a single decoder for convolutional coding or for LDPC coding. This decoder is usually followed by a de-interleaver and a block decoder for some form of BCH block decoding, which is one-dimensional Reed-Solomon (RS) coding in DVB-T and is two-dimensional RS coding in DVB-H. This block decoder corrects errors introduced by impulse noise or short-duration drop-outs in received signal strength, better to avoid impairing the capability of a subsequent decoder for the long-duration TRS coding to overcome protracted drop-outs in received signal strength owing to flat-fading arising from multipath reception at a location near the DTV transmitter.
If the DTV transmitter provides once-repeated transmissions to facilitate iterative-diversity reception, the DTV receiver could include an OFDM demodulator followed by a differential delay network to supply delayed initial transmissions of error-correction-coded (ECC) data to a first turbo decoder concurrently with final re-transmissions of the same ECC data to a second turbo decoder. This would permit parallel iterative operation of two turbo decoders, with exchange of information between them to improve coding gain, much as A. L. R. Limberg proposed doing in 8-VSB.
The parallel iterative operation of two turbo decoders consumes more power than is desirable, particularly in battery-powered receivers. Maximal-ratio code combining is a technique that has been used for combining similar transmissions from a plurality of transmitters in multiple-input/multiple-output (MIMO) networks. Searching for a way to avoid parallel iterative operation of two turbo decoders, A. L. R. Limberg considered the use of maximal ratio code combining of later transmissions of ECC data with earlier transmissions of similar ECC from the same 8-VSB transmitter. The hope was that a combined signal would be generated that could be decoded by iterative operation of a single turbo decoder. One problem encountered when trying to implement such an approach is that the coding of M/H-service data is not independent of the coding of main-service data in 8-VSB broadcasting per the ATSC standard. The inner convolutional coding of the M/H signal is part of a one-half-rate convolutional coding that intersperses main-service signal components with M/H-service signal components. Accordingly, practically considered, the inner convolutional coding of the later transmissions of CCC and the inner convolutional coding of the delayed earlier transmissions of CCC still have to be decoded separately. The outer convolutional coding of the M/H signal is affected by the pre-coding of the most-significant bits of 8-VSB symbols responding to main-service data interspersed among the most-significant bits of 8-VSB symbols responding to M/H-service data. There are also some problems with measuring the energies of the later transmissions of CCC and the delayed earlier transmissions of CCC to provide the information needed for weighting these transmissions for maximal-ratio code combining.
In a replacement system for DTV broadcasting in the United States of America that uses COFDM of a plurality of carrier waves, the FEC coding of main-service data and the FEC coding of M/H-service data can be kept independent of each other. Also, the inclusion of unmodulated carrier waves among the COFDM carrier waves facilitates measurements of their total root-mean-square (RMS) energy in later transmissions and in earlier transmissions of similar data to provide the information needed to weight later and delayed earlier transmissions appropriately for maximal-ratio code combining.
European engineers updated the COFDM transmissions used in the DVB-H standard as originally developed for European broadcasting, so as to support a form of iterative-diversity reception. The orthogonal coordinates of lattice points in 16QAM symbol constellations are rotated, so the imaginary-axis coordinates duplicate the real-axis coordinates. Then the imaginary-axis coordinates of successive 16QAM symbol constellations are delayed a prescribed period of time respective to their real-axis coordinates to provide iterative diversity between the two sets of coordinates. The rotation of the axes of the orthogonal coordinates decreases by a factor of four the spacing between lattice-point coordinates along each axis. The inventor points out that it is preferable to repeat 256QAM symbol constellations without rotation, rather than using rotated 16QAM symbol constellations. The spacing between lattice-point coordinates along each axis is reduced by a factor of four by going from 16QAM symbol constellations to 256QAM symbol constellations, too. The duplication of the 256QAM symbol constellations halves their digital payload. However, sixteen times as many lattice points are available in each 256QAM symbol constellation as in each 16QAM symbol constellation. So, overall, a pair of the repeated 256 QAM symbol constellations provides eight times the digital payload of the rotated 16QAM symbol constellation of same duration as each of the 256 QAM symbol constellations. This eight times larger digital payload can support more forward-error-correction (FEC) coding, if such be desired.
A normal presumption of one skilled in the art of DTV broadcasting casually considering the possibility of repeating frames or time-slices of coded data transmissions is that single-time repetition necessarily will halve digital payload overall. Surprisingly, in actual practice this need not be the case for COFDM DTV transmissions. Customarily, the number of lattice points in the QAM symbol constellation used in modulating the COFDM carrier waves is set by the desired performance in the presence of additive white Gaussian noise (AWGN). Repeating frames or time-slices of COFDM DTV transmissions a single time accommodates the number of lattice points in the QAM symbol constellation used in modulating the COFDM carrier waves being quadrupled, doubling the number of data-slicing bins along each of the I-axis and Q-axis coordinates of each QAM symbol constellation. The smaller size of the data-slicing bins doubles the ratio of AWGN to bin-size for a 6 dB loss of performance in the presence of AWGN. Additively combining the two similar COFDM DTV transmissions improves performance in the presence of AWGN by 3 dB. Alternatively, using code combining techniques can improve performance in the presence of AWGN by somewhat more than 3 dB. The quadrupled number of lattice points in the QAM symbol constellation permits additional FEC coding that can more than recover any remaining portion of the 6 dB loss in performance in the presence of AWGN. The simple convolutional coding (CC) with one-half code rate used in DVB-T can be replaced by parallel concatenated convolutional coding (PCCC) with one-third code rate, for example. When PCCC replaces CC, overall digital payload is two-thirds of what it is in DVB-T, and performance in the presence of AWGN is substantially better than for DVB-T. Puncturing the PCCC to one-half code rate provides the same overall digital payload as DVB-T with possibly as good or even somewhat better performance in the presence of AWGN.
Repeating frames or time-slices of COFDM DTV transmissions a single time may even allow the number of lattice points in the QAM symbol constellation used in modulating the COFDM carrier waves to be increased by a factor of sixteen, quadrupling the number of data-slicing bins along each of the I-axis and Q-axis coordinates of each QAM symbol constellation. The smaller size of the data-slicing bins quadruples the ratio of AWGN to bin-size for a 12 dB loss of performance in the presence of AWGN. Combining similar transmissions can recover 3 DB or more of this loss in performance. The sixteen times increase in the number of lattice points secures eight times more overall digital payload than DVB-T has, even though some of this digital payload must be sacrificed to get as good performance in the presence of AWGN as DVB-T provides.
That is, surprisingly, the reduction in overall code rate that results from repeating COFDM transmissions for iterative-diversity reception can be counteracted by increasing the size of the symbol constellations associated with quadrature amplitude modulation (QAM) of the plural carriers. Increasing the size of the QAM symbol constellations tends to reduce the capability of DTV receivers to decode COFDM transmissions received over the air when accompanied by additive white Gaussian noise (AWGN). FEC coding of data bits is used to facilitate DTV receivers being better able to decode COFDM transmissions accompanied by AWGN. Various types of FEC coding are particularly effective for enabling DTV receivers to overcome AWGN by using iterative decoding procedures called “turbo decoding” because of a fancied resemblance to turbo-charging in automobile engines. The various types of FEC coding that can use turbo decoding procedures are collectively referred to as “turbo coding” in this specification, although the term was originally applied specifically to what is now called parallel concatenated convolutional coding (PCCC). By way of specific examples, turbo decoding procedures are also applicable to serial concatenated convolutional coding (SCCC), to product coding and to parallel concatenated low-density parity-check (LDPC) coding.
Turbo decoding allows QAM symbol constellations larger than 16QAM to be practical in wireless, over-the-air COFDM transmissions of wideband digital signals such as those employed in DTV broadcasting. Signal strength of the COFDM transmissions need not be so large to obtain a large coverage area. So, transmission towers need not be spaced as close together in a single-frequency network (SFN).
A typical receiver for COFDM plural-carrier signals includes an OFDM demodulator, a frequency-domain channel equalizer, and a symbol de-mapper of modulation symbol constellations for successively considered ones of the OFDM carriers. The OFDM demodulator recovers descriptions of the complex amplitude modulation of each carrier, which for digital television broadcasting is usually quadrature amplitude modulation (QAM). The frequency-domain channel equalizer normalizes the descriptions of the amplitudes of the in-phase and quadrature-phase components of each successively considered carrier. The symbol de-mapper converts each successive modulation symbol to a respective set of decision bits. These decision bits may be soft-decision bits, each composed of a hard-decision bit that is either a ONE or a ZERO and of further bits descriptive of the level of confidence that the hard-decision bit is correct.
The hard-decision bits in each successive set of decision bits are associated with one of a number of lattice points in a two-dimensional range of complex amplitude modulation, which number of lattice points define a constellation of ideal complex-amplitude-modulation possibilities. These lattice points correspond to the values of complex amplitude modulation that are transmitted by COFDM plural-carrier signals emanating from the broadcast television station. These lattice points are enumerated consecutively according to some prescribed pattern. Customarily, the number of lattice points in the two-dimensional range of complex amplitude modulation is an integral power of two, and the enumeration employs binary numbers descriptive of respective ones of all the different respective segments of binary coding that are possible.
A receiver for the COFDM plural-carrier signals is apt to recover values of complex amplitude modulation that depart in some degree from lattice points in the two-dimensional range of complex amplitude modulation, owing to imperfect reception. Ongoing departures are caused by Johnson noise arising in the atmosphere and in the receiver elements. Occasional departures are caused by burst noise, often generated by electrical equipment near the receiver. Some departure may arise from imperfect channel equalization filtering. Generally, the further bits of the soft-decision bits associated with the complex amplitude modulation actually received are determined by how far the position defined by that complex amplitude modulation departs from the boundaries of change in the hard-bit values associated with closest lattice point in the two-dimensional range of complex amplitude modulation.
The hard-decision bits in each successive set of decision bits with each lattice point in the two-dimensional range of complex amplitude modulation can be independent of the more significant bits of the in-phase coordinates and quadrature-phase coordinates of the two-dimensional QAM symbol constellation. This permits the commonplace prior-art practice of using Gray mapping of QAM constellations. Paragraphs 0053-0059 of Pat. App. US-2009/0323846-A1 published 13 Nov. 2003 for B. W. Kroeger and titled “Digital audio broadcasting method and apparatus using complementary pattern-mapped convolutional codes” analyzes the Gray mapping of QAM constellations in terms of bit-mapping of in-phase and quadrature-phase amplitude-shift-keying (ASK) components of the QAM symbol constellation. In this mapping procedure the set of decision bits associated with any lattice point differs by only a single bit from the set of decision bits associated with any one of the closest by lattice points in the QAM symbol constellation, so the ASK components exhibit the fewest transitions of bit values possible with any mapping. Kroeger notes in paragraph 0058 of Pat. App. US-2009/0323846-A1 that Gray coding is known to be beneficial upon detection of the ASK signals in noise, since the most likely bit estimation errors are made when the level is near a bit transition. Kroeger observes that the most significant bit of each ASK component exhibits only a single transition in bit value and each successively less significant bit of the ASK component exhibits twice as many transitions in bit value as its predecessor. So, the most significant bit of each ASK component is less susceptible of error than its least significant bit, with varying susceptibilities of error in its other bits. Accordingly, he advocates using the more significant bits of the ASK components for the more important bits of the coding mapped to the QAM symbol constellations and using the more significant bits of the ASK components for the less important bits of the coding mapped to the QAM symbol constellations.
Perfect Gray mapping is possible for a square QAM constellation having an even power of two lattice points therein. I. e., perfect Gray mapping is possible for 16QAM, 64QAM, 256QAM or 1024QAM constellations that are square. The inventor found that perfect Gray mapping cannot be obtained with cruciform QAM constellations having an odd-power-of-two lattice points therein. Conditions at the interior vertices of the cruciform constellation disrupt perfect Gray mapping. I. e., perfect Gray mapping cannot be obtained with cruciform constellations for 8QAM, 32QAM, 128QAM, 512QAM or 2048QAM. However, the inventor has determined that Gray mapping perfect except for double-bit changes between certain sets of decision bits is possible in 8QAM, 32QAM, 128QAM, 512QAM and 2048QAM. Consider the number of lattice points between change in each hard-decision bit within successive sets of decision bits sharing the same in-phase coordinates or the same quadrature-phase coordinates in a two-dimensional QAM symbol constellation. Such numbers vary from hard-decision bit to hard-decision bit within each set of decision bits. The variation in this number for each hard-decision bit exhibits a well-defined pattern if perfect or almost perfect Gray mapping is used.